This invention relates generally to systems and methods for operating a mass flow controller (MFC) and, more specifically, to systems and methods for measuring mass flow within a mass flow controller by sensing the resistance change of a sense resistor or resistors in response to gas flow.
Many manufacturing processes require the flow of process gases to be strictly controlled. To do so, the gas mass flow rate must be sensed and determined. Gas mass flow controllers sense the mass flow rate of a gas substantially independent of gas temperature or pressure, provide a measurement, and meter gas flow to adjust the mass flow rate as desired based on such sensing and metering. Mass flow controllers that operate on heat transfer principles have been widely adapted in the industry.
A common form of mass flow sensor for a gas incorporates a small diameter tube (a capillary tube) having two coils of wire wound on the outside in close proximity to each other, with one coil being positioned upstream of the other. The coils are formed from metallic material having a resistance that is temperature sensitive. The coils are heated by an electrical circuit in a bridge-type electrical circuit incorporated into a sensor to provide equal resistance in the absence of gas flow and hence a balance condition for the bridge-type circuit, i.e., a null output signal.
When gas flows within the capillary tube, the cool incoming gas is warmed as it flows past the upstream element, and this warmer gas then flows past the downstream element, resulting in differential cooling of the two elements. The difference in temperature is proportional to the number of molecules per unit time flowing through the capillary tube. Based on the known variation of the resistance of the coils with temperature, the output signal of the bridge circuit can provide a measure of the gas mass flow.
However, prior art mass flow sensor interface circuits have certain undesirable characteristics. First, prior art circuits compromise the ideal situation where a circuit is simply driving the sensing elements because they include resistances in parallel with the sensing elements for trimming the circuit output to zero volts with zero flow. By doing so, they compromise the apparent gain of the sensor. Sensor gain is traded off for controllability of zero-volt/zero-flow conditions. It is normally undesirable to attenuate the sensor output. If too many impedances of commensurable value are placed in parallel with the sense resistors, they degrade the maximum signal voltage that can be derived from the sense resistors in response to flow.
Secondly, in prior art circuits the relationship between the output voltage (which is proportional to flow) and the difference between the upstream and downstream sense elements is nonlinear, which is an undesirable feature of the prior art. Because the typical resistance values placed in parallel with the upstream and downstream sense elements are not significantly larger than the resistance values of the sense elements, the non-linearity in the relationship between the sense elements and the output voltage is not negligible.
Furthermore, prior art circuits require a large amount of amplification, typically on the order of a 35-70 gain factor, to produce a zero to five-volt output indicative of the gas flow. This requires additional circuitry and complexity. These prior art circuits require that the output from the sensing circuit be connected to a high input impedance amplifier stage because any load placed on the common junction point of the upstream and downstream sense elements increases the loss of sensor output and the circuit non-linearity.
Further, prior art circuits are designed to be calibrated manually using a gain control potentiometers (pots). In addition, the zero-volt/zero-flow condition also is adjusted manually by an operator using a multimeter. This results in increased circuitry and complexity as well as a greater opportunity for inaccurate mass flow values due to drift in the gain and zero control devices (potentiometers).
Prior art circuits are also susceptible to ambient temperature deviations common to both sense elements. Because prior art mass flow sensing circuits are essentially voltage dividers that can be arbitrarily trimmed to zero output by means of a virtual ground and variable resistors, any ambient temperature change will be reflected in the circuit output voltage. This occurs because prior art systems compare the absolute change in the resistance of the sense resistors. In such a case, even if both elements are cooled or heated equally, the absolute resistance change in each will likely be different.
FIG. 1 shows the basic topology of a prior art flow sensing bridge circuit 100. Flow sensing bridge circuit 100 is a modified Wheatstone bridge driven by an ideal current source 20. An ideal current source is characterized by a very high internal impedance, which means its output current will not change with a change in the voltage drop across a load. Ideal current source 20 can thus supply the same current regardless of the voltage drop across the load.
One branch of prior art flow sensing bridge circuit 100 of FIG. 1 consists of two sense elements, RU and RD. These sense elements are used to sense the gas flow and are representative of the respective dynamic resistance values of the upstream and downstream sensor coils wound on the outside of the capillary tube; RU represents the upstream sense element and RD represents the downstream sense element. The upstream sensor coil is cooled more by the gas stream flow than the downstream sensor coil, therefore the resistance value of RU is less than that of RD. With no gas flow, RU is equal to RD and the bridge is balanced by means of variable resistor RV1. Under nonzero flow conditions, prior art flow sensing bridge circuit 100 output voltage eout 30 is given by Equation 1 below. As shown by Equation 1, the relationship between output voltage eout 30 and (RUxe2x88x92RD) is nonlinear.                               e          out                =                                            (                                                                    R                    D                                    ⁢                                      R                    1                                                  -                                                      R                    U                                    ⁢                                      R                    2                                                              )                        ⁢            i                                                              (                                                      R                    U                                    +                                      R                    D                                                  )                            ⁢                              (                                  1                  +                                                                                    R                        1                                            +                                              R                        2                                                                                    R                      p                                                                      )                                      +                          (                                                R                  1                                +                                  R                  2                                            )                                                          [EQN.  1]            
The other branch of flow sensing bridge circuit 100 includes resistors R8, R9 and variable resistor RV1. The impedance value of variable resistor RV1 is only a small fraction of the values of resistors R8 and R9. Variable resistor RV1 is used to adjust the offset of flow sensing bridge circuit 100 so that output voltage eout 30 is zero. Resistance value R1 represents the combined value of resistor R8 and the portion of variable resistor RV1 on the resistor R8 side of variable resistor RV1""s wiper arm, and resistance value R2 represents the combined value of resistor R9 and the portion of variable resistor RV1 on the resistor R9 side of variable resistor RV1""s wiper arm. Typically, the values of R1 and R2 are about eight times as large as those of sense resistors RU and RD. Additionally, resistor Rp is connected in parallel with sense resistors RU and RD and resistors R8 and R9. The value of Rp is about four times as large as that of R1 and R2. The non-linearity of the circuit is therefore non-negligible.
As can be seen in Equation 1 above, the relationship between output voltage eout 30 and sense element resistances RU and RD is inherently non-linear. Additionally, Resistors RV1, R8, R9 and Rp connected in parallel with sense elements RU and RD reduce the effect of sense elements RU and RD. Ideally, the entire circuit current should run through sense elements RU and RD to obtain the maximum output signal. FIG. 1 therefore demonstrates both the non-linear characteristics of the prior art as well as the reduction in output voltage eout 30 resulting from the use of resistances in parallel with sense elements RU and RD.
FIG. 2 is a more detailed representation of the prior art flow sensing bridge circuit 100 of FIG. 1. Individual components of ideal current source 20 are shown in detail. The amplitude of the current applied to flow sensing bridge circuit 100 is controlled by a voltage derived from reference voltage source (+5 Vref) 32 applied across resistor R100. The wiper of variable resistor RV1 provides a virtual ground controlled by means of operational amplifier U3 and transistor Q2. Capacitor C1 provides stabilizing feedback (lag compensation). The bridge output voltage is filtered by resistor R11 and capacitor C2. The filter comprised of resistor R11 and capacitor C2 provides a xe2x88x923 DB cut-off frequency approximately equal to 600 Hz. eout 30 is the filtered output voltage.
+5V reference voltage 32 is inverted by means of operational amplifier U1 to become xe2x88x925 volt. The xe2x88x925 volt output of operational amplifier U1 is compared to the voltage drop across resistor R200. Most notably, the xe2x88x925 volt output of operational amplifier U1 is scaled down to become +1.2775 volt nominal at the output of operational amplifier U2. Although the output of operational amplifier U2 is not exactly 1.2775 volts, it causes transistor Q1 to control the current through reference resistor Rr and the rest of flow sensing bridge circuit 100 in such a fashion that the voltage drop across resistor R9 equals the scaled down value of the xe2x88x925 volt reference (which has been scaled down and inverted by operational amplifier U2). Therefore, the drop across reference resistor R8, which can be a 100 kilo-ohm resistor, is a precision 1.2775 volt. Consequently, the current through reference resistor Rr is 12.775 mA nominal. The remaining resistors in FIG. 2 are all 1% resistors: R3, R4, R5, R6, R7, R10, R12, and R14, as well as the other resistors previously mentioned.
Ideal current source 20 provides a constant current source regardless of the load being driven by the circuit. This is because the feedback control circuit comprised of operational amplifier U2, transistor Q1, current reference resistor Rr and scaling components R3, R4, R6 and R7, maintains the voltage drop across current reference resistor Rr at 1.2775 volt nominal. As long as the circuit parameters remain within the linear operating range of component values, the value of the current through current reference resistor Rr will remain a precisely controlled and constant value.
When any impedance is driven by an ideal current source such as that in FIG. 2 and the resulting output signal is to be used for further processing, another voltage reference is needed for the signal processing amplifier. The voltage reference in the prior art circuit is provided by variable resistor RV1. The voltage reference can either be actual or virtual. A circuit as shown in FIG. 1 without the grounding at the wiper arm of variable resistor RV1 would represent a totally undefined circuit.
The circuit comprising operational amplifier U3 and transistor Q2 produces a virtual ground at variable resistor RV1. The virtual ground is produced by comparing the voltage at the wiper of RV1 to ground potential at the positive input node of operational amplifier U3. The output of operational amplifier U3 drives transistor Q2 and is controlled in such a fashion as to reduce to zero the voltage between the positive and the negative nodes of operational amplifier U3.
Operational amplifiers typically have an open loop gain factor of anywhere from ten thousand times to a million times, referred to as the differential node voltage. Ideally, the wiper potential at variable resistor RV1 is maintained at zero volts. The deviation from zero is small so as to be negligible. Flow sensing bridge circuit 100 voltage is thus referenced to zero volts.
The output of flow sensing bridge circuit 100 is taken at the junction of upstream and downstream flow sense resistors RU and RD. When flow sensing bridge circuit 100 is balanced at zero flow by means of variable resistor RV1, the zero voltage that exists at the wiper of variable resistor RV1 is mirrored over to the junction of sense elements RU and RD; this feature is an inherent property of a Wheatstone bridge. However, when the balance is offset because the resistance value of upstream sense element RU becomes smaller than the resistance value of downstream sense element RD, while the other branch of flow sensing bridge circuit 100 maintains its previous set balance, the deviation from the balanced state will manifest itself as an output voltage, eout 30.
Referring back to FIG. 1, some typical impedance values are included for R1 and R2, which are individually about 4.07 kxcexa9. Resistors R8 and R9 are individually 4.02 kxcexa9 resistors and variable resistor RV1 is 100 xcexa9. If the circuit is perfectly balanced and equal, 50 xcexa9 of RV1 are apportioned to R8 and 50 xcexa9 to R9, hence the 4.07 kxcexa9 value for R1 and R2. RP is typically about 17.4 kxcexa9 and sense elements RU and RD individually are about 500 xcexa9 at working temperature.
It is to be realized that one purpose of the current flowing through sense elements RU and RD is to heat them up so that gas flow can differentially cool sense elements RU and RD, thus sensing gas flow. However, impedances R1 and R2 are only about eight times the value of RU and RD, hence their shunting effect is non-negligible, and in fact is quite large. The shunting effect is a limitation of prior art circuits because it reduces the value of output voltage eout 30.
A need exists for an improved mass flow interface circuit that provides the capability to increase sensor output voltage and eliminate the prior art problems associated with output signal reduction by providing a much smaller shunting effect. This can be accomplished by using very large resistances connected in parallel with the upstream and downstream sense elements, and thereby increase the overall effectiveness of the ideal current source.
A further need exists for an improved mass flow sensor interface circuit that eliminates the non-linearity existing in prior art mass flow sensor interface circuits. By using much larger resistances in parallel with the sense elements, the non-linearity effect can be greatly reduced from the prior art circuit.
An even further need exists for an improved mass flow sensor interface circuit that eliminates the need to manually calibrate the circuit using zero and gain adjustments, thereby eliminating the drift problems associated with the prior art circuitry due to temperature and also reducing the sensitivity to vibration which is more likely to affect the adjustable resistors.
Similarly, a still further need exists for an improved mass flow sensor interface circuit that is inherently insensitive to ambient temperature changes and thereby eliminates the temperature drift problems associated with prior art circuits. This is especially advantageous if the circuitry is being operated in an enclosed case where ambient temperature is likely to change.
An even further need exists for an improved mass flow sensor interface circuit that does not require a large degree of amplification of the circuit output voltage to produce a signal indicative of gas flow, and that thus eliminates the need to further process the output voltage signal through an amplifier with a much larger input impedance than the bridge impedances.
In accordance with the present invention, a mass flow sensor interface circuit is provided that substantially eliminates or reduces the disadvantages and problems associated with previously developed mass flow sensor interface circuits. In particular, the present invention provides an improved mass flow sensor interface circuit and method for sensing and measuring mass flow rate of a gas to provide an output voltage proportional to the gas flow rate in a mass flow controller.
The improved mass flow sensor interface circuit of the present invention includes an upstream mass flow sense element, a downstream mass flow sense element, and a precision current source to drive the circuit. The circuit further includes and operational amplifier stage to sum the voltage upstream of the upstream sense element with the voltage downstream of the downstream sense element. A reference voltage is electrically connected to the positive node of the operational amplifier. An upstream shunting resistor and a downstream shunting resistor share a common junction at the negative node of the operational amplifier and are electrically connected in parallel to the upstream and downstream mass flow sense elements.
The present invention further includes a reference resistor electrically connected between the reference voltage source and the positive node of the operational amplifier and a feedback resistor electrically connected between the output and the positive node of the operational amplifier. An output voltage proportional to the resistance change across the upstream and downstream sense elements, and therefore proportional to the gas mass flow rate, is provided as an output of the operational amplifier. The improved mass flow sensor interface circuit of the present invention provides a more linearized output, greater sensitivity, greater reliability and better accuracy than prior art such circuits.
The voltages upstream of the upstream sense element and downstream of the downstream sense element are referenced to +2.5 volts and are added by the operational amplifier. The node voltages of the operational amplifier are maintained at +2.5 volts, with the negative node serving as a virtual ground.
The sensitivity of the mass flow sensor interface circuit of the present invention can be greatly increased over that of such prior art circuits. The effect of the shunting resistors placed in parallel with the upstream and downstream sense elements on circuit sensitivity is minimal. The value of the shunting resistors can typically be about 100 times that of the sensing elements with the result that the non-linear response caused by the summation of the impedance values of the upstream and downstream sense elements in prior art circuits is substantially reduced. The gain of the improved circuit of the present invention can be increased by the appropriate choice of the feedback resistor of the operational amplifier.
The precision current source is further comprised of a series of resistors and operational amplifiers designed to maintain a constant current through the upstream and downstream sense elements. Furthermore, a signal conditioning amplifier can be used to condition the output signal derived from the resistance change in the upstream and downstream sense elements before feeding the output signal to an analog-to-digital converter.
The present invention provides an important technical advantage for an improved mass flow interface circuit that provides the capability to increase sensor output voltage and eliminate the prior art problems associated with output signal reduction by providing a much smaller shunting effect through the use of very large resistances connected in parallel with the upstream and downstream sense elements, and thereby increase the overall effectiveness of the ideal current source.
Yet another technical advantage of the improved mass flow sensor interface circuit of the present invention eliminates the non-linearity existing in prior art mass flow sensor interface circuits. By using much larger resistances in parallel with the sense resistors, the non-linearity effect is greatly reduced over prior art circuits. The ideal current source driving the circuit is made more ideal because the sense resistors see a higher impedance from the current source.
Still another technical advantage of the improved mass flow sensor interface circuit of the present invention eliminates the need to manually calibrate the circuit using potentiometers for the zero flow/zero trim position, thereby eliminating the drift problems associated with the prior art circuitry due to temperature and bumping or jarring of the circuit, which could cause potentiometer misalignment.
A still further technical advantage of the improved mass flow sensor interface circuit of the present invention is that it is inherently insensitive to ambient temperature changes and thereby eliminates the temperature drift problems associated with prior art circuits. This is especially advantageous if the circuitry is being operated in an enclosed area where ambient temperature is likely to change.
An even further technical advantage of the improved mass flow sensor interface circuit of the present invention is that it does not require a large degree of amplification of the circuit output voltage to produce a signal indicative of gas flow, thus eliminating the need to further process the output voltage signal through an amplifier with a much larger input impedance than the bridge impedances.